Rf detector with crest factor measurement

ABSTRACT

An RF detector configured to provide two outputs, one being a function of the true RMS power level of an RF input signal, and the other being a function of the instantaneous/peak power of the RF input signal, normalized to the average power level. The RF detector includes a variable gain detection subsystem including a single detector or detector array that provides a representation of the power level of the RF input signal. The detector or detector array is common to both the RMS power detection channel and the instantaneous/peak power detection channel of the RF detector. A method of RF detection includes providing representations of the RF input signal at different gain levels, selecting one or more of the representations, and averaging the selected signals. The gain levels of the selected representations is adjusted to provide information about the average power level of the RF input signal.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a continuation of, and claims priority under 35U.S.C. §120 to, co-pending U.S. application Ser. No. 12/152,634 entitled“RF DETECTOR WITH CREST FACTOR MEASUREMENT,” filed May 14, 2008, whichclaims priority under 35 U.S.C. §119(e) to U.S. Provisional applicationNo. 60/930,120 filed May 14, 2007 and entitled “RF DETECTOR,” each ofwhich is herein incorporated by reference in its entirety.

BACKGROUND

1. Field of Invention

The present invention relates generally to RF detectors and, moreparticularly, to RF detectors capable of providing an indication of boththe average power and the instantaneous or peak power levels of an RFsignal.

2. Discussion of Related Art

There are many applications in which it is desirable to measure the peakand/or average power level of a radio frequency (RF) signal. Forexample, power measurement and control of the RF signals in both thetransmitting and receiving chains of modern wireless communicationssystems, such as cellular telephone networks, may be essential. Toefficiently use the available bandwidth, the transmitted signals inthese systems may be modulated using complex modulation standards suchas CDMA, WCDMA or WiMax. These complex modulated signals have a timevarying crest factor, which is defined as the peak to average powerratio of the signal, resulting in intolerable errors if conventionalpower detectors using diode detection or successive amplification areused to measure signal power.

Conventional techniques to characterize modulated signals depend on theparallel processing of the input signal to compute the average power andthe instantaneous or peak power. For example, referring to FIG. 1A, asingle RF input signal at terminal 102 is processed by an envelopedetector or peak detector 104 to generate the instantaneous power/peakpower output, a, and is also processed by an average power detector 106to generate the average power output, b. FIG. 1B illustrates a variationof this method in which the RF input signal is initially processed usinga power envelope detector 112 and is then processed using an averagingcircuit 114 to generate the average power output, b, and using thebuffer/peak detector 104 to generate the instantaneous/peak poweroutput, a.

In some cases, it is desirable to measure the crest factor of the RFsignal and/or to obtain information about the signal wave shape.Calculation of the crest factor requires both average power informationand peak power information. Referring again to FIG. 1A, a conventionaltechnique uses a divider 108 to calculate the crest factor afterparallel processing of the RF input signal to determine the peak power,a, and the average power, b, as discussed, for example, in U.S. Pat. No.5,220,276. The divider 108 calculates the ratio of the peak detector 104output (a) to the average power output, b, resulting in an output signal(b/a) on terminal 110 that is a representation of the input signal crestfactor. When an envelope detector is used instead of a peak detector,the divider provides an instantaneous power output signal that isnormalized to average power; i.e., the ratio of instantaneous power toaverage power.

A disadvantage of RF detectors using the parallel processing techniqueis that an RF coupler (not shown) is required at the input 102 to driveboth the average power detection channel and the envelope power or peakpower detection channel. In addition, because different detectors (104and 106) are used in the two different detection channels, there mayexist part-to-part, process, and temperature variations between the twochannels, which can degrade the accuracy of the measurements,particularly of the crest factor measurement. Matching circuitry may berequired to compensate for such differences between the two channels,adding complexity and cost to the system. Another disadvantage is thatthe divider 108 may be required to handle nonlinear calculationsdepending on the output characteristics of the average power detectionchannel and the envelope power or peak power detection channel. Inaddition, because any inaccuracy in the divider would compromise thecrest factor measurement, an accurate, possibly complex and expensive,divider may be required.

There are a variety of commercially available average power detectorsthat may be used in the systems illustrated in FIGS. 1A and 1B. Oneexample of an average power detector is an RMS-DC converter. RMS-DCconverters are used to convert the RMS (square-root of the mean(average) of the square) value of an input AC (time-varying) signal intoa DC (or quasi-DC) current or voltage. RMS-DC converters are capable ofmeasuring the RMS power of an RF signal independent of the signal waveshape or crest factor. Wide-dynamic range average power detectors usingfeedback control loop techniques are commercially available.

For example, referring to FIG. 2, there is illustrated a block diagramof an RMS-DC converter 200 that incorporates a squaring RF detector 206with a variable scale factor and a feedback control loop. The RMS-DCconverter receives an RF input, Vin, at terminal 202 and provides asignal at terminal 204 which is representative of the average power ofthe input signal. The squaring RF detector 206 is responsive to ascaling factor control signal, Vscale, received at a control port 208and provides an output voltage, Vout, at an output port 209, the outputvoltage being a representation of the square of the RF input signalscaled by a monotonic function of the control signal. Thus, the outputof the squaring RF detector 206 is given by:

V _(out) =|V _(in)|² ×f(V _(scale))  (1)

The second element of the average power detection feedback loop is anintegrator 210 having an input port 212 coupled to the output port 209of the squaring RF detector 206, a reference port 213 that receives areference signal 214, and an output port 216 coupled to the control port208 of the squaring RF detector 206. The output port 216 of theintegrator 210 is also coupled to the terminal 204 of the RMS-DCconverter 200. The integrator 210 is configured to integrate thedifference between the output, Vout, of the squaring RF detector 206 andthe reference signal 214 to adjust the scaling factor of the squaring RFdetector until the average output signal of the squaring RF detector isequal to the reference signal, thus resulting in a feedback controlloop. This feedback loop forces the squaring RF detector 206 to operateat a controlled output operating point. For example, a drop in the RFinput signal power received at terminal 202 results in negativeintegration in the integrator 210, forcing the squaring RF detector 206to provide amplification to the input signal to keep the averagesquaring RF detector output, Vout, at a constant point. Because of thisinteraction in the feedback control loop, the scaling factor controlsignal, Vscale, of the squaring RF detector 206 will vary as a functionof the average of the RF input signal, Vin, providing a representationof this RF input signal average power. Some examples of such, orsimilar, RMS-DC converters are disclosed in U.S. Pat. Nos. 6,348,829,6,429,720 and 6,437,630.

Single-detector average power or peak power detecting schemes, such asthose illustrated in FIGS. 1A and 1B, suffer from a reduced dynamicrange, for example, on the order of about 35 dB in high-frequencyintegrated circuit detector designs. Average power detector using afeedback control loop technique, such as that illustrated in FIG. 2 anddiscussed above can achieve much larger dynamic range depending on thescaling function implementation in the feedback control loop. Forexample, average power detectors with more than 75 dB dynamic range arecommercially available. However, in a system such as that illustrated inFIG. 1A, where both peak/envelope and average power are measured, thenormalized instantaneous power output or peak power output would belimited by the lesser performance (dynamic range) of the envelope poweror peak power detecting scheme. In addition, single-detector envelopepower or peak power detecting scheme are generally also highly dependenton the input RF frequency, which may not be desirable in manyapplications.

SUMMARY OF INVENTION

Aspects and embodiments of the present invention are directed to awide-dynamic range RF detector that accepts a modulated or unmodulatedRF input signal and provides an output which varies as a quasi-linearfunction of the logarithm of the RMS value of the RF input signalvoltage. That is, the RF detector provides an output that varieslinearly (or nearly so) with the RMS voltage measured in dB of the RFinput signal. The RF detector is also capable of providing a secondoutput representative of the instantaneous or peak power level, of theRF input signal, normalized to the average power of the signal.Embodiments of the RF detector use a single detector array for bothaverage (e.g., RMS) power detection and instantaneous/peak powerdetection, thereby eliminating some of the part-to-part, process and/ortemperature variability issues which may exist with conventionalsystems, as discussed above. In addition, embodiments of the RF detectorimplement a feedback control loop to increase the input dynamic rangeand normalize the measured instantaneous/peak power to the averagepower, thereby removing the need for an accurate divider. Thus, asdiscussed further below, the RF detector circuit according to aspectsand embodiments of the invention may be used to provide accurateindications of the average power level, normalized instantaneous powerlevel, and peak-to-average power ratio (crest factor) of an RF signalover a wide range of input power levels and modulation complexity.

According to one embodiment, a power detector comprises an inputconfigured to receive an input signal, a variable gain detectionsubsystem coupled to the input and that detects the input signal andprovides a detector output signal, an integrator coupled to the variablegain detection subsystem and configured to receive the detector outputsignal and a reference signal and to provide an integrator output signalwhich is representative of an average power level of the input signal,and an instantaneous power processing device coupled to the detectorsubsystem and configured to receive the detector output signal and toprovide at an output of the power detector an instantaneous power outputsignal which is representative of the instantaneous power level of theinput signal normalized to the average power level of the input signal,wherein the variable gain detection subsystem is configured to receivethe integrator output signal and to adjust the detector output signal toa level approximately that of the reference signal.

In one example of the power detector, the variable gain detectionsubsystem comprises at least one squaring detector. The variable gaindetection subsystem may further include a variable gain amplifiercoupled between the input and the at least one squaring detector,wherein the variable gain amplifier is configured to receive theintegrator output signal and to provide an amplified output signal,wherein the variable gain amplifier is configured so that the gain ofthe variable gain amplifier is controlled by the integrator outputsignal, and wherein the squaring detector is configured to receive theamplified output signal. In another example, the variable gain detectionsubsystem includes a gain stage configured to provide a plurality ofgain tap signals from the input signal, a plurality of detectorsconfigured to receive the plurality of gain tap signals and to provide acorresponding plurality of detector tap signals, and means for selectingat least one of the plurality of detector tap signals responsive to acontrol signal and providing the at least one selected detector tapsignal as the detector output signal, wherein the control signal is afunction of the integrator output signal.

In another example, the variable gain detection subsystem comprises afirst series of gain stages coupled in series and configured to providea first plurality of gain tap signals, a second series of gain stagescoupled in series and configured to provide a second plurality of gaintap signals, and a plurality of multipliers coupled to the first andsecond series of gain stages, wherein each multiplier is configured tomultiply a respective one of the first plurality of gain tap signalswith a respective one of the second plurality of gain tap signals toprovide a plurality of squared signals. In another example, the variablegain detection subsystem includes a gain stage configured to provide aplurality of gain tap signals from the input signal, a plurality ofdetectors configured to receive a the plurality of gain tap signals andto provide a corresponding plurality of detector tap signals, and aninterpolator coupled between the plurality of detectors and theintegrator and between plurality of detectors and the instantaneouspower processing device, wherein the interpolator is configured toreceive the plurality of detector tap signals, to select at least onedetector tap signal, and to provide an interpolator output signal thatis a function of the at least one selected detector tap signal, andwherein the detector output signal received by the integrator and theinstantaneous power processing device comprises the interpolator outputsignal. The interpolator may be configured to select the detector tapsignals from at least those detectors operating in their square-lawregion. The interpolator output signal may be a function of a weightedsum of the selected detector tap signals. In another example, the gainstage comprises a plurality of amplifiers coupled in series andconfigured to amplify the input signal to provide a plurality ofamplified signals; and wherein the plurality of gain tap signalscomprises the plurality of amplified signals. The gain stage may alsoinclude a plurality of attenuators configured to attenuate the inputsignal to provide a plurality of attenuated signals, and the pluralityof gain tap signals may comprise the plurality of amplified signals andthe plurality of attenuated signals. The interpolator may comprise aplurality of interpolator stages, each interpolator stage configured toreceive a respective one of the plurality of detector tap signals, anindividual fixed bias reference signal and a common control signal,wherein the common control signal is derived from the integrator outputsignal. In one example, each interpolator stage comprises a controllablecurrent amplifier, and wherein a gain of the controllable currentamplifier is a function of the individual fixed bias reference signaland the common control signal.

In one example of the power detector, the instantaneous power processingdevice includes a low-pass filter that filters the detector outputsignal. The low pass filter may have a time constant which is smallerthan an output time constant of the integrator. In another example, theinstantaneous power processing device includes an amplifier to amplifythe detector output signal. In a further example, the instantaneouspower processing device comprises a peak detector, and the peak detectoroutput signal is representative of the peak power level of the inputsignal normalized to the average power level of the input signal. Theinstantaneous power processing device may also comprise a comparatorconfigured to receive the detector output signal and the referencesignal and to generate an error signal based on a subtraction betweenthe detector output signal and the reference signal, and an output blockconfigured to receive the error signal and to provide the instantaneouspower output signal. In one example, the output block comprises anoutput buffer, which may be implemented a transistor follower. Inanother example, the output block comprises a peak detector thatprovides a peak detector output signal, wherein the peak detector outputsignal is representative of the peak power level of the input signalnormalized to the average power level of the input signal. The peakdetector may be implemented using a transistor follower coupled to acapacitor, wherein the capacitor stores a voltage representative of apeak signal level at an output of the peak detector.

Another example of the power detector comprises an at least partialreplica instantaneous power processing device configured to provide a DCreference bias output signal, and a summer configured to receive the DCreference bias signal and the instantaneous power output signal and tosubtract the DC reference bias signal from the instantaneous poweroutput signal to generate a DC offset-adjusted instantaneous poweroutput signal, wherein a signal provided at the output of the powerdetector comprises the adjusted instantaneous power output signal whichis representative of the instantaneous power level of the input signalnormalized to the average power level of the input signal. In anotherexample, the instantaneous power processing device comprises a peakdetector, and the signal provided at the output of the power detectorcomprises an adjusted peak detector output signal which isrepresentative of the peak power level of the input signal normalized tothe average power level of the input signal.

Another aspect is directed to a method of power detection comprisingreceiving an input signal, detecting a power level of the input signalto provide a detected signal, comparing the detected signal with areference signal to provide an error signal, integrating the errorsignal to provide an integrator output signal which is representative ofan average power level of the input signal, providing an instantaneouspower output signal responsive to the detected signal, the instantaneouspower output signal being representative of the instantaneous powerlevel of the input signal normalized to the average power level of theinput signal, and adjusting the detected signal to a level approximatelythat of the reference signal. In one example, detecting the level of theinput signal comprises squaring the input signal. The method may furthercomprise amplifying the input signal with a variable gain amplifier toprovide an amplified signal, and adjusting a gain of the variable gainamplifier responsive to the integrator output signal, wherein detectinga level the input signal includes detecting a level of the amplifiedsignal. In one example, providing the instantaneous power output signalincludes detecting a peak of the detected signal and providing a peakpower output signal normalized to the average power level of the inputsignal. In another example, providing the instantaneous power outputsignal includes comparing the detected signal with a second referencesignal to provide a second error signal, filtering and buffering thesecond error signal to provide the instantaneous power output signal.

According to another aspect, a method of power detection comprisesproviding a plurality of representations of an input signal at differentgain levels, detecting the plurality of representations of the inputsignal to provide a corresponding plurality of detected signals,selecting at least one of the detected signals to provide at least oneselected signal, averaging the at least one selected signal to providean averaged signal, providing an integrator output signal representativeof an average power level of the input signal based on the averagedsignal, and providing an instantaneous power output signalrepresentative of an instantaneous power level of the input signalnormalized to the average power level of the input signal based on theat least one selected signal. In one example, providing a plurality ofrepresentations of the input signal further includes attenuating theinput signal to provide a plurality of attenuated signals; and whereinthe plurality of representations of the input signal further includesthe plurality of attenuated signals. In another example, selecting atleast one of the detected signals includes interpolating between theplurality of detected signals to provide at least two selected signals,weighting the at least two selected signals to provide at least twoweighted signals, and summing the at least two weighted signals toprovide an interpolator output signal. In another example, selecting atleast one of the detected signals includes selecting those detectorsignals from detectors operating in their square law region. In oneexample, detecting the plurality of representations of the input signalincludes squaring the plurality of representations of the input signal,and providing the corresponding plurality of detected signals includesproviding a corresponding plurality of squared signals. In anotherexample, providing the instantaneous power output signal may includedetecting a peak of the detected signal and providing a peak poweroutput signal normalized to the average power level of the input signal.

According to another aspect, a method of power detection comprisesgenerating a series of gain tap signals from an input signal, squaringand weighting each of the gain tap signals, thereby generating a seriesof weighted output signals, summing the weighted output signals, therebygenerating a summed output signal, providing an integrator output signalrepresentative of an average power level of the input signal based onthe summed output signal, and providing, responsive to the summed outputsignal, an instantaneous power output signal representative of aninstantaneous power level of the input signal normalized to the averagepower level of the input signal.

Still other aspects, embodiments, and advantages of these exemplaryaspects and embodiments, are discussed in detail below. Moreover, it isto be understood that both the foregoing information and the followingdetailed description are merely illustrative examples of various aspectsand embodiments, and are intended to provide an overview or frameworkfor understanding the nature and character of the claimed aspects andembodiments. The accompanying drawings are included to provideillustration and a further understanding of the various aspects andembodiments, and are incorporated in and constitute a part of thisspecification. The drawings, together with the remainder of thespecification, serve to explain principles and operations of thedescribed and claimed aspects and embodiments.

BRIEF DESCRIPTION OF THE DRAWINGS

Various aspects of at least one embodiment are discussed below withreference to the accompanying figures. In the figures, which are notintended to be drawn to scale, each identical or nearly identicalcomponent that is illustrated in various figures is represented by alike numeral. For purposes of clarity, not every component may belabeled in every drawing. The figures are provided for the purposes ofillustration and explanation and are not intended as a definition of thelimits of the invention. In the figures:

FIG. 1A is a block diagram of one example of a conventional system formeasuring the crest factor of an input signal;

FIG. 1B is a block diagram of another example of a conventional detectorsystem;

FIG. 2 is a block diagram of one example of a conventional RMS-DCconverter;

FIG. 3 is a block diagram of an RF detector according to aspects of theinvention;

FIG. 4 a block diagram of another example of arrangement of componentsin the instantaneous/peak power detection channel of an embodiment ofthe RF detector according to aspects of the invention;

FIG. 5 is a block diagram of one example of an RF detector according toaspects of the invention;

FIG. 6 is a block diagram of another example of an RF detector accordingto aspects of the invention;

FIG. 7 is a flow diagram illustrating one example of a method of RFdetection according to aspects of the invention;

FIG. 8 is a block diagram of another example of an RF detector accordingto aspects of the invention;

FIG. 9 is a block diagram of one example of a variable gain detectionsubsystem, according to aspects of the invention;

FIG. 10 is a block diagram of another example of an RF detectoraccording to aspects of the invention;

FIGS. 11A and B together are a block diagram of another example of aportion of an RF detector including an example of an interpolator,according to aspects of the invention;

FIG. 12 is a diagram of one example of an interpolator according toaspects of the invention;

FIG. 13 is a block diagram of another example of an RF detectoraccording to aspects of the invention;

FIG. 14 is a circuit diagram of one example the instantaneous powerprocessing block and an output buffer according to aspects of theinvention;

FIG. 15A is an illustration of an example of an output signal providedat the instantaneous power output of one embodiment of the RF detectoraccording to aspects of the invention;

FIG. 15B is an illustration of an example of an RF input signalcorresponding to the output signal illustrated in FIG. 15A;

FIG. 16 is a block diagram of another example of an RF detectoraccording to aspects of the invention;

FIG. 17 is a circuit diagram of an example of the instantaneous powerprocessing block and an output peak detector according to aspects of theinvention;

FIG. 18 is a block diagram of an example of a portion of a variable gaindetection subsystem according to aspects of the invention;

FIG. 19 is a block diagram of another example of an RF detectoraccording to aspects of the invention; and

FIG. 20 is a block diagram of another example of an RF detectoraccording to aspects of the invention.

DETAILED DESCRIPTION

There is a wide range of applications in which providing an accuratemeasurement or representation of both the true RMS (root-mean-square)power and an indication of the instantaneous or peak power of an RFsignal may be advantageous. For example, in communications systemssignal power level measurements may be used to provide a received signalstrength indication (RSSI) and/or a transmitter signal strengthindication (TSSI). These signal power measurements may also be used forRF power amplifier efficiency control, receiver automatic gain control,and/or transmitter power control. As discussed above, some systemsinvolve complex modulated signals (e.g., CDMA, WCDMA or WiMAX wirelesscommunication systems). These systems may benefit from accurate averagepower information that is independent of the modulation scheme. In somecommunication systems, instantaneous or peak power level information, incombination with RMS average power level information may be critical toavoid saturating components in the signal processing chain. Someadaptive power amplifier biasing techniques may also require, or benefitfrom, crest factor knowledge to accurately set the power amplifieroperating conditions for efficient power versus linearity tradeoffs.

Accordingly, aspects and embodiments are directed to a wide-dynamicrange RF detector that is capable of providing an indication of theinstantaneous or peak power level, normalized to the average powerlevel, of an RF input signal in addition to the true RMS power level. Asdiscussed above, feedback loop control may be used for wide-dynamicrange average power detection. The RF detector may provide an accurateindication of the average real power of an RF input signal, independentof the signal shape or crest factor, and thus independent of themodulation scheme applied to the RF input signal. In addition, the RFdetector may provide normalized instantaneous power detection thatmirrors the input RF modulation envelope, as discussed further below.

It is to be appreciated that embodiments of the methods and apparatusesdiscussed herein are not limited in application to the details ofconstruction and the arrangement of devices set forth in the followingdescription or illustrated in the accompanying drawings. The methods andapparatuses are capable of implementation in other embodiments and ofbeing practiced or of being carried out in various ways. Examples ofspecific implementations are provided herein for illustrative purposesonly and are not intended to be limiting. In particular, acts, elementsand features discussed in connection with any one or more embodimentsare not intended to be excluded from a similar role in any otherembodiments. Also, the phraseology and terminology used herein is forthe purpose of description and should not be regarded as limiting. Theuse herein of “including,” “comprising,” “having,” “containing,”“involving,” and variations thereof is meant to encompass the itemslisted thereafter and equivalents thereof as well as additional items.

Referring to FIG. 3, there is illustrated a block diagram of one exampleof an RF detector according to aspects of the invention. The RF detectorreceives an RF input signal at an RF input terminal 302. This inputsignal is processed by a variable gain detection subsystem 304 thatprovides a power detector output signal on line 306. The scaling factorbetween the level of the detector output signal on line 306 and the RFinput power level is controlled by a scaling factor control signalsupplied on line 318, as discussed further below. It is to beappreciated that the variable gain detection subsystem 304 may beimplemented in a variety of ways, at least some of which are discussedin more detail below. For example, In one embodiment, the variable gaindetection subsystem may include a squaring RF detector cell (or array ofsuch cells) along with circuitry to adjust the scaling factor of thedetector cell(s), as shown in FIG. 5. In another embodiment, thevariable gain detector subsystem 304 may include a variable gainamplifier 328 preceding the squaring RF detector cell(s) 330, as shownin FIG. 6. In another embodiment, the variable gain detection subsystemmay provide a plurality of representations of the power level of the RFinput signal at different gain levels and these representations may beprocessed by a detection block and scaled (either during or subsequentto the detection) and summed to provide the detector output signal online 306, as discussed further below.

According to one embodiment, the detector output signal on line 306 maybe provided to an average power detection path which provides an output308 representative of the true mean-square or root-mean-square (RMS)level of the RF input signal applied at the RF input terminal 302. Aslave copy of the detector output signal on line 306 is provided to aninstantaneous or peak power detection path which provides an output 310representative of the instantaneous or peak power level of the RF inputsignal, as also discussed further below. The output 308 is referred toherein as the RMS output 308, and the signal provided thereat isreferred to as the RMS output signal. However, it is to be appreciatedthat the signal provided at output 308 may be a representation of themean-square level of the RF input rather than the RMS level. Similarly,the output 310 is referred to herein as the instantaneous power output310, and the signal provided thereat is referred to as the instantaneouspower signal. However, it is to be appreciated that in some examples thesignal provided at the output 310 may be representative of the peak(rather than instantaneous) power level of the RF input signal, asdiscussed further below. The instantaneous power output 310 isnormalized to the mean power of the RF signal. In one embodiment, theinstantaneous power output 310 follows any amplitude modulation of theRF input signal in such a way that the signal swing on the instantaneouspower output varies with the instantaneous power of the modulated RFinput signal, as discussed further below.

Still referring to FIG. 3, the average power detection path includes anintegrator 312 that averages the detector output signal received on line306, thus providing at the RMS output 308 a signal that isrepresentative of the average power of the RF input signal applied atthe RF input terminal 302. In one embodiment, the average powerdetection path includes a comparator 314 that compares the signalreceived from the variable gain detection subsystem 304 (on line 306) toa fixed reference signal, Iref1, provided by a reference generator 346(see FIG. 8). The comparator may be implemented in a variety of ways, asknown to those skilled in the at, including, but not limited to, asubtractor that generates an error signal by subtracting the receiveddetector output signal from the reference signal, or vice versa. Theintegrator 312 may be responsive to the error signal provided on line316 from the comparator 314. The integrator may be implemented as ananalog circuit, in which case its output will vary in continuousfashion, or as a digital accumulator, in which case its output wouldvary in discrete steps as a stream of binary data and either or both theRMS output 308 and line 318 on which the scaling control signal issupplied may represent digital busses. It is also to be appreciated thatalthough the comparator 314 is illustrated in FIG. 3 as a separatecomponent from the integrator 312, the comparator 314 may be integratedwith the integrator 312 into a single component block and the integrator312 may be responsive to both the detector output signal (on line 306)and the reference signal, Iref1. The RMS output signal may be providedon line 318 to a control input of the variable gain detection subsystem304 to adjust the scaling factor of the variable gain detectionsubsystem, as discussed further below. Thus, the integrator 312(including the comparator 314) forms a feedback control loop with thevariable gain detection subsystem 304 to adjust the scaling factor ofthe variable gain detection subsystem until the average detector outputsignal on line 306 is approximately equal to the reference signal,Iref1, as discussed further below. The scaling factor control of thevariable gain detection subsystem 304 is a monotonic function of thesignal provided on line 318, such as a linear function, a squaringfunction, or an exponential function. This function may be implementedin continuous (analog) fashion or in discrete steps in which case, asdiscussed above, line 318 may represent a digital control bus

As illustrated in FIG. 3, the instantaneous power detection pathincludes an instantaneous power processing block 320 and an output block322. As discussed above, in one embodiment, a slave copy of the detectoroutput signal on line 306 from the variable gain detection subsystem 304is fed to the instantaneous power processing block 320, the output ofwhich provided, via the output block 322, at the instantaneous poweroutput 310. Because of the feedback control loop discussed above, theaverage value of the detector output signal on line 306 is equivalent tothe reference signal, Iref1, resulting in the detector output signalrepresenting the instantaneous power of the input RF signal normalizedto average power of the RF input signal. The slave copy of this signalcan be buffered, scaled and filtered (in any order) by the instantaneouspower processing block 320 and output block 322 to generate theinstantaneous power output normalized to average power.

Still referring to FIG. 3, the instantaneous power processing block 320may be configured to scale and/or filter and/or level-shift the detectoroutput signal. In one example, the instantaneous power processing blockis configured to low-pass filter the detector output signal received online 306. Filtering may be useful to remove RF frequency ripple that mayexist at the output of the variable gain detection subsystem 304 on line306 while preserving the RF input signal modulation envelope. Theinstantaneous power processing block may be further responsive to asecond reference signal, Iref2, provided by the reference generator 346(see FIG. 8), and may be configured to compare the detector outputsignal received on line 306 to the reference signal Iref2 using acomparator 324. Operation of the comparator 324 may be considered alevel-shifting operation on the detector output signal. The output block322 is responsive to the output of the instantaneous power processingblock 320. In one example, the output block 322 includes an outputbuffer, wherein the output of the instantaneous power processing block320 is buffered with the output buffer to generate the output 310 thatis representative of the instantaneous power level of the RF inputsignal normalized to average RF input signal power, as discussed furtherbelow. In another example, the output block 322 includes a peak detectorthat may be used to generate a representation of the peak RF inputsignal power normalized to average RF input signal power (also definedas the crest factor of the RF input signal), as also discussed furtherbelow.

As noted above, buffering, scaling and/or filtering of the detectoroutput signal may be accomplished in any order. For example, asillustrated in FIG. 4, the detector output signal from line 306 may befiltered, and optionally amplified or gain-scaled, by block 326 of theinstantaneous power processing block 320, and then compared with thereference signal, Iref2, in comparator 324 before being buffered byoutput block 322. In addition, it is to be appreciated that the value ofthe reference signal, Iref2, applied to the comparator 324 may be thesame as or different to the value of the reference signal, Iref1, usedin the average power detection channel. Furthermore, in some examples,Iref1 may be fed to the instantaneous power processing block 320 (aswell as to the first integrator 312) rather than generating a separatesecond signal, Iref2.

Referring again to FIG. 3, as discussed above, the output block 322 mayinclude a peak detector that may be used to generate the peak poweroutput normalized to the average power, and thus to provide ameasurement of the crest factor of the RF input signal. Similar toinstantaneous power detection configuration, filtering maybe useful toremove the RF frequency ripple existing at the output of the variablegain detection subsystem 304, while preserving the input modulationenvelope and improving crest factor accuracy. As discussed above,conventional systems that provide a crest factor measurement, and thusrequire both an average power measurement and a peak power measurement,implement parallel processing of the RF input signal and may thereforesuffer disadvantages such as part-to-part variations, temperaturevariations, and different dynamic range capability over the twodetection channels, as well as the need for an accurate divider tocalculate the crest factor. An RF detector such as that illustrated inFIG. 3 (or variations thereof) may overcome challenges associated withconvention RF detectors by dominantly using the components in awide-dynamic range average power detector having a feedback control loopwith minimal additional complexity. The addition of theinstantaneous/peak power output 310 may be implemented using only aslave copy of the detector output signal (which is used in the averagepower detection channel); optionally including some gain/filteringcomponents, and a buffer and/or peak detector, as discussed above.Furthermore, the slave copy of the detector output signal may onlyoperate up to the input signal modulation bandwidth frequency, forexample, about 40 MHz, instead of the input RF frequency, which maysimplify the components required for the instantaneous power processingblock 320 and/or output block 322.

As discussed above, the RF detector may be implemented using variousembodiments of the variable gain detection subsystem block 304. Forexample, referring to FIG. 6, in one embodiment, the RF input signal atinput terminal 302 may be fed to a variable gain amplifier 328. Thefeedback signal on line 318 from the output of the integrator 312 may beused to control the gain of the variable gain amplifier. By sweepingthrough the gain range of the variable gain amplifier 328 by adjustingthe feedback signal on line 318, representations of the RF input signalmay be obtained at sequentially varying levels of gain. The output ofthe variable gain amplifier 328 may drive a squaring detector 330 whichprovides the detector output signal on line 306. As discussed above,because of the feedback control loop, the average value of the detectoroutput signal on line 306 is driven to approximately match the referencesignal, Iref1, and therefore, the system may reach a steady state whenthe RMS output signal (which is fed back on line 318) selects a gainlevel of the variable gain amplifier 328 that results in the detectoroutput signal on line 306 being approximately equal to the referencesignal, Iref1. It is to be appreciated that of course gain orlevel-shifting may be provided on line 306 (not shown) such that thedetector output signal is driven to a level that is different from thereference signal, Iref1, by a factor of the gain or level-shiftingapplied on line 306.

As also illustrated in FIG. 6, in another embodiment, the RF detectormay include additional squaring detectors 332 which may be used togenerate the reference signals, Iref1 and Iref2. In this manner, processand/or temperature variations of the squaring detector 330 may becancelled in the comparison (or subtraction) operations of thecomparators 314 and 324.

In another embodiment, the variable gain detection subsystem 304includes a squaring detector 330 having a variable squaring gain that iscontrolled by a bias control circuit 331, as illustrated in FIG. 5. TheRF input signal at input terminal 302 is fed to the squaring detector330 which provides the detector output signal on line 306, as discussedabove. The squaring detector 330 may include one or more square lawdetector cells. In this embodiment, the scale control feedback signal online 318 drives the bias control circuit 331 which varies the squaringgain of the squaring detector 330 by adjusting the bias current throughthe square-law detector cell(s) included in the squaring detector 330.The detector output signal on line 306 is used to drive the scalecontrol feedback loop in the same way as discussed above with referenceto FIGS. 3 and 6. In one example, the bias control circuit 331 may alsoadjust the Iref1 and Iref2 bias currents through the additionalreference squaring detectors 332 shown in FIG. 6.

The RF detector may be used to implement various different methods of RFdetection. A flow diagram of one example of a method of RF detection isillustrated in FIG. 7. Aspects and embodiments of the method of RFdetection are discussed below with continuing reference to FIG. 7.

In one embodiment, a first step 502 includes providing representationsof an RF input signal at different gain levels. As discussed above, anRF signal input to the RF detector at the RF input terminal 302 may beprocessed by the variable gain detection subsystem 304 which maygenerate multiple representations of the RF input signal at differentgain levels.

According to one embodiment, the variable gain detection subsystem 304of the RF detector includes a gain stage 334 which includes a chain ofamplifiers 336 which amplify the RF input signal, and a chain ofattenuators 338 which attenuate the RF input signal, as illustrated inFIG. 8. It is to be appreciated that the invention is not limited to theinclusion of both amplifiers and attenuators, nor to inclusion of anequal number of amplifiers and attenuators. Rather, embodiments of theRF detector may comprise any number of, and any combination of,amplifiers and/or attenuators. By applying the input RF signal at inputterminal 302 to the chains of amplifiers 336 and attenuators 338, aplurality of taps of the input RF signal can be generated that areseparated from one another by specific amounts of gain. These taps arereferred to herein as “gain taps” of the RF input signal. It is to beappreciated that the term “gain” as used herein refers to both positivegain, for example, as provided by an amplifier, and “negative gain” asprovided, for example, by an attenuator. Thus, referring back to FIG. 7,step 502 may include amplifying the RF input signal to provide gain tapsat different amplified levels (step 504), and attenuating the RF inputsignal to provide gain taps at different attenuated levels (step 506).

Referring again to FIG. 8, according to one embodiment, the variablegain detection subsystem 304 also comprises a detector array 340 thatincludes a plurality of detectors 342. In one example, a detector 342 isprovided for each amplifier in the chain of amplifiers 336 and for eachattenuator in the chain of attenuators 338 to detect a signal level atthe output thereof. The outputs from the detectors 342 may be fed to aninterpolator 344 which is configured to select the outputs of those oneor more detectors operating in, or closest to, an optimum square lawregion, as discussed further below. The interpolator 344 provides thedetector output current, Iout, on line 306 which is a function of thecombined outputs from the selected detectors 342. The position of theselected outputs in the chain of amplifiers 336 and/or chain ofattenuators 338 in the gain and detector subsystem 304 may provideinformation that is representative of the average power level of the RFinput, as discussed further below. The interpolator may also include acontrol input 348 which receives the feedback signal on line 318 fromthe RMS output 308, as discussed further below.

Referring to FIG. 9, there is illustrated a block diagram of one exampleof the variable gain detection subsystem 304. As discussed above, thevariable gain detection subsystem 304 includes a gain stage 334 whichmay include a chain of amplifiers 336 and a chain of attenuators 338. Inthe illustrated example, the chain of attenuators 338 is implemented asa chain of resistors. However, it is to be appreciated that theinvention is not so limited and other attenuator elements may be used,as known to those skilled in the art. In one example, each amplifier inthe chain of amplifiers 336 may have the same gain, for example, X dB.Similarly, each attenuator in the chain of attenuators 338 may providethe same level of attenuation, for example, also X dB of attenuation (or−X dB of gain). In this example, the RF input signal is fed seriallythrough each amplifier and attenuator, as illustrated in FIG. 9.Assuming that there are N amplifiers in the chain of amplifiers 336,each having a gain of X dB, and M attenuators in the chain ofattenuators 338, each having a gain of −X dB, then including a neutralpoint, there are (M+N+1) gain taps sequentially separated by X dB.Alternatively, the amplifiers and/or attenuators may have differentgains and, depending on the gain values and the manner in which the RFinput signal to applied to each amplifier and/or attenuator, a number ofgain taps may be generated that are sequentially separated by the sameor different amounts of gain.

For example, referring to FIG. 10, there is illustrated another exampleof an RF detector in which the gain stage 334 comprises a plurality ofamplifiers 336 k each having a gain that is a multiple of X dB (e.g.,aX, bX, cX, . . . nX). The gain coefficients (a, b, c . . . n) may beinteger or non-integer coefficients. As discussed above, the gain stage334 may comprise any combination of amplifiers and/or attenuators, andtherefore the example illustrated in FIG. 10 is not intended to belimiting and the gain stage 334 may additionally (or alternatively)include a number of attenuators each having a gain that is a multiple of−X dB. As illustrated in FIG. 10, in this example, the RF input signalis applied to each amplifier 336 k in parallel, rather than in series(as in the example illustrated in FIG. 13). Thus, if there are Namplifiers 336 k and if the gain coefficients (a, b, c . . . n) aresequential integers (e.g., 1, 2, 3, etc.), then the gain stage 334 willgenerate N gain taps sequentially separated by X dB.

In one embodiment, each of the gain elements (amplifiers and/orattenuators) in the gain stage 334 is implemented differentially.However, it is to be appreciated that the invention is not so limitedand single-ended implementation is also possible. Referring again toFIG. 9, in the illustrated example, each gain tap is buffered (usingbuffers 350) before driving the detector array 340. Tap separation maybe stabilized over temperature and supply and process variations usingappropriate biasing techniques. It is to be appreciated that althoughthe buffers 350 are illustrated in FIG. 9 as being included with thechain of amplifiers 336 and chain of attenuators 338, the invention isnot so limited and the buffers 350 may instead be considered as existingsubsequent to the gain stage 334.

As illustrated in FIG. 7, a next step 508 may include detecting each ofthe gain taps produced by the gain stage 334. According to oneembodiment, each buffered gain tap from the chain of amplifiers 336 andchain of attenuators 338 drives a detector 342 which provides a(differential or single-ended) output current. Thus, the detector array340 produces (M+N+1) current outputs which are fed to the interpolator344. Similarly, in the example illustrated in FIG. 10, the detectorarray 340 produces N current outputs which are fed to the interpolator344. It is to be appreciated that although various inputs and outputs ofthe RF detector and its components may be described herein as currentsor voltages, the invention is not so limited and any of these inputsand/or outputs may be easily converted from a voltage to a current, orvice versa, using techniques known to those skilled in the art. In oneexample, the detectors 342 are squaring detectors, and the outputcurrent signal from each detector may ideally vary as the square of theinput voltage (i.e., the input gain tap). Thus, the step 508 ofdetecting the gain taps may include squaring the gain taps. The outputs352 from the detector array 340 may be filtered by filters 354 prior tobeing fed to the interpolator 344, as illustrated in FIG. 9. In oneexample, the filters 354 may be low-pass filters and used to reduce RFfrequency ripple on the signals.

As will be recognized by those skilled in the art, the detectors 342 maybe implemented in various different ways, and may have fixed or variablescale factors, as discussed further below. In one embodiment, eachdetector 342 may be implemented as a common-emitter triplet cell, whichmay be chosen to have a high transistor ratio for extending the inputvoltage range over which a square-law output current may be obtained inpractice. Similarly, the variable gain detection subsystem 334 may beimplemented in various different ways, as also discussed further below.

Referring again to FIG. 7, the method may include, after the detectingstep 508, a step 510 of selecting one or more of the detected(optionally squared) representations of the RF input signal. All thedetectors 342 are driven by the gain taps corresponding to differentsignal levels, as discussed above, but only a small number of thedetectors may be operating within their optimum square-law region. Largesignals at the input of a detector 342 may drive the detector into itslimiting region rather than the square-law region. Input signals thatare too small may be masked by voltage offsets and noise that may bepresent in practical embodiments. Therefore, as discussed above, in oneembodiment, the interpolator 344 is configured to select the outputcurrents from those detectors 342 that are operating in their optimumsquare-law region. Selection may be performed using a series of fixedreference voltages and a single control voltage, V_(cont), which isderived from the output of the integrator 312 and provided to theinterpolator 344 at the control input 348, as discussed further below.Smooth, well-controlled interpolation of the outputs from adjacentdetectors 342 as a function of the control voltage, V_(cont), may bedesirable for accurate logarithmic RMS signal level indication.

Referring to FIG. 11, there is illustrated a block diagram of an exampleof the RF detector, showing an example of the interpolator 344 in moredetail. As discussed above, the interpolator 344 receives the(optionally filtered) outputs 352 from the detector array 340 of thevariable gain detection subsystem 304 and is configured to select one ormore of those outputs to be summed and output on line 306 to theintegrator 312 and instantaneous power processing block 320. Accordingto one embodiment, the interpolator 344 comprises a series ofinterpolator stages 356, one stage for each detector output. Thus, ifthe detector array 340 provides (M+N+1) outputs 352, the interpolator344 may comprise (M+N+1) interpolator stages 356. In one example, eachinterpolator stage 356 has three inputs and receives the output of adetector (at input 358), the control voltage, Vcont, (at control input360), and a fixed bias reference voltage (at bias input 362), asdiscussed further below. Also as discussed further below, the controlvoltage, Vcont, may be derived from the scaled or unsealed output of theintegrator 312. The interpolator 344 may include (or be coupled to) abias generator 364 which generates the series of fixed bias referencevoltages applied to the interpolator stages 356, as also discussedfurther below. The outputs from all the interpolator stages 356 may becombined to provide the interpolator output current, Iout, (alsoreferred to as the detector output signal) on line 306.

According to one embodiment, the interpolator stages 356 are implementedas controllable current amplifiers with the current outputs connectedtogether, as illustrated in FIG. 12. Each interpolator stage 356 mayreceive the current from one of the detector outputs at the input 358. Aspecific amount of this received current may be transferred to theoutput line 306 (as Iout), and the control input 360 (which receives thecontrol voltage, Vcont) and bias input 362 of the interpolator stage 356may be used to set the current gain of the interpolator stage. Asillustrated in FIG. 12, in one example, all the control inputs 360 ofthe interpolator stages 356 are connected in common to the controlvoltage input terminal 348 of the interpolator 344. In one embodiment,when the control voltage, Vcont, is close to the bias reference voltage,Vref, for a given interpolator stage 356, the current gain for thatstage may be maximum, and the gain may drop sharply when Vcont is eithersmaller or larger than Vref. Thus, the current gain of each interpolatorstage weights the current from the respective detector 342. In oneexample, each interpolator stage 356 is configured to provide an outputonly when the control voltage is within a predetermined level of thebias reference voltage of that stage. For better accuracy of the overalllogarithmic RMS power level indication (provided at the RMS output 308),the gain of each interpolator stage 356 as a function of the controlvoltage, Vcont, may be chosen not to be symmetrical around Vref.

As discussed above, the outputs from all the interpolator stages 356 aresummed to provide the detector output signal, Iout. Each individualinterpolator stage 356 has a current gain determined by the differencebetween the control voltage, Vcont, and the fixed bias reference voltagefor the individual, Vrefi. Thus, those detector output currents 352 thatare applied to interpolator stages 356 with relatively high currentgains are selected by the interpolator 344. The currents from theselected detectors 342 are weighted by the current gain of therespective interpolator stages 356 and thus, the interpolator outputcurrent, Iout, may be considered a weighted sum of the outputs fromselected detectors 342.

Referring again to FIGS. 11 and 12, in one example, the bias inputs 362of the interpolator stages 356 are driven by buffered fixed referencevoltages that are generated by the bias reference generator 364.Buffering may be provided by buffers 366 connected between the biasreference generator 364 and each bias input 362, as illustrated in FIG.12. These fixed reference voltages may be separated from each other byan amount ΔV, with the lowest reference voltage having a value Vofs.Thus, there may be (M+N+1) reference voltages, Vrefi, starting from Vofsand increasing sequentially by ΔV up to Vofs+(M+N)*ΔV. Of course, it isto be appreciated that the sequence of reference voltages, Vrefi, may bereversed, such that the first reference voltage, Vref1, corresponds tothe highest voltage (Vofs+(M+N)*ΔV) and values of the reference voltagesdecrease sequentially by the amount ΔV, such that Vref(M+N+1) is thelowest voltage, equal to Vofs. In this example, the lowest RF inputsignal requiring the largest gain may be represented by the lowestvoltage at the RMS output 308. The absolute accuracies of thesereference voltages, Vrefi, may directly affect the accuracy of thesignals provided at the RMS output 308 and instantaneous power output310. If the reference voltage separation ΔV is selected with care, whenthe control voltage, Vcont, is swept, the output of the interpolator 344(i.e., the combined current, Iout, from all the interpolator stages 356which is provided on line 306) may vary smoothly between K times theminimum detector tap current and K times the maximum detector tapcurrent, where K is the scaling amount of the interpolator.

According to one embodiment, the interpolator 344 may comprise a secondtype of interpolator stage 356 b that is different from other stages 356that may receive a current from the detector 342 that is driven by thehighest gain tap of the gain stage 334. In one embodiment, as shown inFIG. 12, when the control voltage, Vcont, is close to the fixed biasreference voltage, Vref, for this interpolator stage 356 b, the currentgain of the interpolator stage 356 b may be close to the maximum and thegain may drop sharply when Vcont is larger than Vref while the gain willapproach its maximum value when Vcont is smaller than Vref. In anotherembodiment, when the control voltage, Vcont, is close to the fixed biasreference voltage, Vref, for this interpolator stage 356 b, the currentgain of the interpolator stage 356 b may be close to the maximum and thegain may drop sharply when Vcont is smaller than Vref while the gainwill approach its maximum value when Vcont is larger than Vref. In thismanner, the output of the interpolator 344 (which is provided on line306) changes monotonically from minimum to maximum as a function of theVcont level rather than tending towards a minimum value at both thehighest and lowest levels of Vcont.

According to another embodiment, the detector array 340 and interpolator344 may be designed to be temperature stable when used together in anembodiment of the RF detector. Temperature stability may be achievedusing precise biasing circuits with specific temperaturecharacteristics, as known to those skilled in the art.

Referring again to FIGS. 8 and 9, in another “thermometer” interpolatorembodiment of the variable gain detection subsystem 304, currents fromdetectors 342 driven by gain/attenuator chain taps with signal levelsabove the optimum square-law region of the detector are suppressed, butthe currents from those detectors 342 operating in their optimumsquare-law region are summed with the small, offset-driven currents fromthose detectors 342 connected to tap points with signals below optimum.These offset-induced currents may diminish the accuracy of thisinterpolation scheme, but this embodiment may allow the instantaneouspower output 310 to provide logarithmic peak amplitude measurement ofshort RF power level spikes well beyond the square-law dynamic range ofthe individual detectors 342. This may be a useful feature in somefault- or interference-monitoring applications. In one embodiment of the“thermometer” interpolator, the interpolator stages 356 in FIG. 12 maybe replaced by stage type 356 b. The current gain of all theseinterpolator stages would then be high when the control voltage is belowthe fixed bias reference voltage for a given stage, i, (that is, forVcont<Vrefi) and drop sharply as the control voltage goes above the biasreference voltage (i.e., for Vcont>Vrefi).

Referring again to FIG. 11, the control voltage, Vcont, applied to theinterpolator stages 356 may be derived from the scaled or unsealedoutput of the integrator 312, which in turn receives, on line 306, theoutput from the interpolator 344. Thus, the RF detector may comprise afeedback loop that includes the interpolator 344 and the integrator 312.In one embodiment, the integrator 312 comprises the comparator 314 thatdetects a difference between the detector output current, Iout, suppliedon line 306 and the reference current, Iref1, provided by the referencegenerator 346. In one example, the comparator 314 is a currentcomparator. However, as discussed above, it is to be appreciated thatany signal referred to herein as a current may be easily replaced with acorresponding voltage and therefore, the comparator may receive anoutput voltage from the interpolator 344 and a reference voltage and maybe a voltage comparator.

Still referring to FIG. 11, the comparator 314 may generate an errorcurrent, Ierror, based on the difference between Iout and Iref1. Thiserror current may be used to charge or discharge the integrationcapacitance, depending on the direction of the error.

The integration capacitance may include one or more capacitors 368, 370.Additional capacitance may be added to the integrator 312 if higherintegration time constants are desired or needed to get a constant RMSpower level indication as a function of the modulation bandwidth of theRF input signal. The integrator 312 averages the signals from thedetectors 342 selected by the interpolator 344 and provided to theintegrator 312 on line 306 (step 512 in FIG. 7). An output amplifier 372may be used for buffering, gain-scaling and/or DC-offset adjustment ofthe signal that is provided at the RMS output 308. In one example, theoutput amplifier 372 may provide suitable current buffering and outputvoltage swing capability to drive an external load that may be connectedto the RMS power output 308. In one example, a loop amplifier 374 isused to provide further scaling and/or buffering of the integratoroutput as may be needed or desired to drive the control input 348 of theinterpolator 344. Furthermore, it is to be appreciated that eitheramplifier 372 and/or amplifier 374 may be implemented as a cascade oftwo or more amplifiers, as would be recognized by those skilled in theart. Thus, a feedback control loop is formed involving the interpolator344, the integrator 312 and the amplifiers (gain/scale buffers) 372 and374.

According to one embodiment, the integrator 312 has a large gain(although not an infinite gain, due to circuit non-idealities that wouldexist in practical embodiments). This large gain in the feedback loopforces the detector output current, Iout, to be the same (or nearly thesame) as the integrator reference current, Iref1, as discussed above.When an RF signal is applied to the input 302 (see FIG. 8) of the RFdetector, or when the RMS level of the applied RF input signal changes,the integrator 312 adjusts the control voltage, Vcont, applied to theinterpolator 344 and receives the corresponding weighted sum of thecurrents from all the detectors 342 selected by the interpolator. Thefeedback loop may reach a steady state condition when the controlvoltage, Vcont, drives the interpolator 344 to select those detectors342 with a weighted average output approximately equal to the referencecurrent, Iref1.

In one example, the interpolator 344 selects the outputs of thedetectors 342 which are operating at (or close to) the optimum squaringpoint, that is, those detectors 342 which provide an accuraterepresentation of the square of the RF input signal voltage. Asdiscussed above, this representation of the square of the RF inputsignal voltage may be obtained regardless of the wave shape of the RFinput signal. In one example, when this result is achieved by selectinga correct scaling (provided by a combination of the interpolator 344 andthe amplifiers 372, 374) and reference current, Iref1, value, themajority of the interpolator output current, Iout, may be supplied byone or two detectors 342. Thus, by selecting from all the detectors 342provided in the RF detector only those detectors 342 operating in theirsquare-law region for a given RF input signal power level, a measurementof the mean-square power level of the RF input signal may be obtainedover a range of RF input signal power levels that may far exceed thesquare-law (dynamic) range of any single detector 342. The interpolator344 may thus select the detector(s) whose output provide the broadestpeak to average signal power range for accurate square law operationover a wide range of RF input power levels.

According to one embodiment, when the output of the integrator 312stabilizes to its steady-state condition, the value of the controlvoltage, Vcont, determines which detector outputs 352 are selected bythe interpolator 344. Thus, by the design of the RF detector, themean-square value of the gain taps to these detectors 342 is known. Inone example, an X dB change in the signal level of the RF input signal,X dB also being the gain/attenuation tap ratio (as discussed above),forces the interpolator-integrator feedback loop to select a consecutiveinterpolator stage 356. Since this may correspond to a shift of ΔV atthe control input 348 of the interpolator 344 (as discussed above), thecontrol voltage, Vcont, may change linearly for logarithmic change inthe mean-square voltage level of the RF input signal. The root operationto obtain the RMS output signal from the mean-square signal level may beimplemented through gain-scaling, for example, by the amplifier 372. Thelogarithm of the root of the mean-square value of the RF input signal isequal to half the logarithm of the mean-square value of the RF inputsignal. Thus, at the RMS output 308, the RF detector may provide anaccurate representation of the RMS power level of the RF input signalover a wide dynamic range (step 514). An additional summing andamplifier circuit (not shown in FIG. 8) may optionally be connected tooutput 308 to act as a buffer preventing an external load fromdisturbing the integrator-interpolator feedback loop. The summingcircuit allows an external power level control signal to be subtractedfrom the signal at output 308, thereby allowing the RF power detector tooperate as a controller in a power leveling loop.

In one embodiment, the accuracy of the RF detector may depend on thestability of amplifier/attenuator gain in the variable gain detectionsubsystem (304), and on the absolute value of the fixed referencevoltages at the bias inputs 362 of the interpolator stages 356. Withknowledge of how much gain is available from the gain stage 334, theexact logarithmic RMS value of the RF input signal may be absolutelyprovided by the control voltage, Vcont, with appropriate offsetcompensation and/or gain scaling from the amplifier 372.

Referring to FIG. 13, there is illustrated another example of an RFdetector including the RMS output 308 and the instantaneous power output310. As discussed above, the gain stage 334 may be implemented in avariety of ways and FIG. 13 illustrates another variation in which thegain stage 334 comprises a plurality of serially connected amplifiers336. However, it is to be appreciated that any of the illustratedvariations (or any of numerous other variations) of the gain block 334may be used in any embodiment of the RF detector.

Accurate RMS power calculation for signals with complex modulationschemes, such as discussed above, may require a long integration time toinclude the time-varying envelope in the measurement. Therefore, theintegration time constant associated with the integrator 312 may bechosen to be relatively long. By contrast, a low-pass filtering timeconstant for the instantaneous power processing block 320 (in theinstantaneous power detection channel) may be chosen so as to suppressfact AC current fluctuations that may be induced by the RF carriersignal while also being sufficiently short so as to follow slowercurrent variations that may be driven by amplitude changes in thecarrier modulation envelope. In one example, the instantaneous powerprocessing block 320 provides an output current, and this output currentmay be easily transformed into a voltage using a simple resistornetwork, as known to those skilled in the art. This transformed voltageoutput of the instantaneous power processing block 320 may provide anindication of the instantaneous amplitude modulation on the RF inputsignal. Furthermore, as discussed above, since it is being generated bythe gain tap point(s) selected by the RMS power detection loop, thevoltage amplitude will be scaled by the RMS power of the RF carrier.Accordingly, the voltage signal from the instantaneous power processingblock 320 may provide a direct indication of the instantaneous toaverage power ratio of the RF input signal modulation over a wide rangeof average RF power levels. Furthermore, in one embodiment, theinstantaneous power level indication, provided from the instantaneouspower processing block 320 at the instantaneous power output 310 (step516), will be accurate for large modulation crest factors because theinstantaneous power processing block 320 is being driven by detectors342 operating in their optimum square-law region. In another embodiment,with the “thermometer” interpolation scheme discussed above, theinstantaneous power level indication range may be extended to provide alogarithmic-peak power indication for modulation crest factors manyorders of magnitude larger than the square-law dynamic range of theindividual detectors 342.

Referring to FIG. 14, there is illustrated one example of animplementation of the instantaneous power processing block 320 andoutput block 322, illustrated collectively as block 392. As discussedabove, the instantaneous power processing block 320 receives a copy ofthe detector output signal from line 306 on its input line 378. In oneexample, the instantaneous power processing block 320 includes, or iscoupled to, the comparator 324 (see FIG. 4) which receives the detectoroutput signal on line 306 and compares it to a reference current, Iref2,(e.g., by subtracting the reference current from the detector outputsignal or vice versa) to generate an error signal that is input on line378 to drive the stage of the instantaneous power processing block 320illustrated in FIG. 14. Therefore, it is to be appreciated that thesignal supplied on line 378 may be the detector output signal from line306 (optionally scaled or level-shifted) or the error current generatedby comparator 324. As discussed above, in one example, the referencecurrent, Iref2, may be equal to the reference current, Iref1, that issubtracted from the detector output signal by comparator 314 in theaverage power detection channel. In this example, the input current online 378 of the instantaneous power processing block 320 will be nearlyzero when the RF input signal at input terminal 302 is not amplitudemodulated.

According to one embodiment, the instantaneous power processing block320 performs a current-to-voltage conversion on the input signal on line378 in the network comprising resistors 382 and 384 and the transistors386 a, 386 b and 386 c. In one example, these transistors 386 a, 386 band 386 c are BJT transistors, as illustrated in FIG. 14, but it is tobe appreciated that they may be alternatively implemented as FETtransistors. The resulting voltage on line 376 is supplied to the outputblock 322. As discussed above, in one embodiment, the signal provided atthe instantaneous power output 310 is a representation of theinstantaneous power of the input RF signal normalized to the mean powerof the input RF signal. In this embodiment, the output block 322comprises a transistor 388 in a follower configuration to buffer thevoltage received on line 376. The transistor 388 is properly biasedusing the resistor 402 which may be internal, external or partiallyexternal (for example, if multiple physical resistors are used toimplement the representative resistor 402) to the output block 322. Inthe illustrated embodiment in FIG. 14, the configuration of transistors386 a, 386 b and 386 c and the resistors 382 and 384 cancels thetemperature variation of the base-emitter voltage of the followertransistor 388 resulting in a temperature stable (or nearly temperaturestable) instantaneous power output signal at the instantaneous poweroutput 310.

Referring to FIG. 15A, there is illustrated a plot of one example of asignal provided at the instantaneous power output 310 of an embodimentof the RF detector. In the example shown in FIG. 15A, the RF inputsignal applied at terminal 302 was a 4-tone signal with a crest factorof 9.04 dB as illustrated in FIG. 15B. As discussed above, and as can beseen in FIG. 15A, in this example, the voltage provided at theinstantaneous power output follows the amplitude modulation shape of theRF input signal, normalized to the average power level of the RF inputsignal.

In one embodiment, a replica 394 of the block 392 may be used as areference to compensate for temperature, process and supply variations,as illustrated in FIG. 16. In the illustrated example, the replica block394 does not receive any input signal and thus provides an outputvoltage that is equivalent to the output voltage of the block 392 whenthere is no input current received on line 378. Accordingly, the signalprovided from the replica block 394 on the output line 398 will be a DClevel that varies with temperature, supply and process variations.Because the replica block 394 is a copy of the block 392, temperature,process and supply variation related errors in the instantaneous poweroutput signal on line 310 may be cancelled by the subtraction of thereference output signal on line 398 using summer 400. As discussedabove, in one example where Iref1 is approximately equal to Iref2, thedifference of the instantaneous power output signal on line 310 andinstantaneous power reference signal on line 398 may be nearly zero ifthe input RF signal does not have any amplitude modulation. Accordingly,the difference signal provided at output 396 may provide arepresentation of the instantaneous power of the input RF signalnormalized to the mean power of the input RF signal with the average ofthis difference signal being near zero. In another embodiment, if Iref2is also applied to port 378 of replica block 394, the difference of theinstantaneous power output signal on line 310 and reference signal online 398 will correspond to a fixed value representative of thereference level for the average power normalization. Accordingly, thedifference signal provided at output 396 may provide a representation ofthe instantaneous power of the input RF signal normalized to the meanpower of the input RF signal with the average of this difference signalbeing at the reference level for the average power normalization.

As discussed above, in another embodiment, the signal provided at theinstantaneous power output 310 may be representative of the peak powerlevel of the RF input signal, normalized to the average power of the RFinput signal. Accordingly, the output block 322 may include a peakdetector that can be used to generate the peak power output normalizedto the average power, also defined as the crest factor of the RF inputsignal. Referring to FIG. 17, there is illustrated another example ofthe instantaneous power processing block 320 and output block 322 inwhich the output block 322 includes a peak detector. As discussed abovewith reference to FIG. 14, this block 392 performs a current-to-voltageconversion on the signal received on line 378 using the networkcomprising resistors 382 and 384 and transistors 386 a, 386 b and 386 c.FIG. 14 illustrates these transistors as BJTs, but it is to beappreciated that they may be alternatively implemented as FETtransistors. In one example, the transistor 388, with its base receivingthe resulting voltage on line 376, is used in a follower configurationwith a large biasing resistor 402 and a capacitor 404 connected to theemitter of the transistor 388. The biasing resistor 402 in this case ismuch larger than the corresponding resistor 402 in FIG. 14 buffer stageconfiguration, so the quiescent current through transistor 388 is muchsmaller. As a result, this transistor charges-up capacitor 404 duringpeaks in the input RF signal power, but it turns off (acts as arectifier) as the RF power drops below the peak value. Thus voltagecorresponding to the peak RF power level is maintained on capacitor 404with a time constant related to the product of the values of resistor402 and capacitor 404. Both resistor 402 and capacitor 404 may beinternal or external, or any combination thereof, to the output block322. In the configuration illustrated in FIG. 17, the block 392 providesat the instantaneous power output 310 a peak power output signal that isa representation of the peak power of the input RF signal normalized tothe mean power of the input RF signal. The ratio of the peak power tomean power is defined as the crest factor.

Thus, in one embodiment, a signal representative of the crest factor ofthe RF input signal may be provided at the instantaneous power output310. In this embodiment, the normalization (or “division) of the peakpower by the average power occurs without the need for an accuratedivider, as is needed in the conventional system discussed above withreference to FIG. 1A, which may greatly simplify the circuit and/orimprove the accuracy of the crest factor measurement. Furthermore, inone embodiment, because the same detector array in the variable gaindetection subsystem 304 is used for both the RMS power measurement andinstantaneous power measurement, part-to-part and temperature variationissues between the RMS power detection channel and the instantaneouspower detection channel, which can be problematic in conventionalsystems, as discussed above, are eliminated. The use of the samedetector array also provides similar dynamic range performance for boththe average power detection function and the instantaneous powerdetection function. In addition, the RF detector circuitry may besimplified compared to conventional systems by the sharing ofcomponents, such as the detector array, among the two detectionchannels.

Referring again to FIG. 16, it is to be appreciated that the block 392and replica block 394 may be implemented as illustrated in FIG. 17.Thus, in similar fashion to the instantaneous power configurationdiscussed previously, the difference of the peak power output signal online 310 and peak power reference output signal on line 398 may provideat output line 396 a representation of the peak power of the input RFsignal normalized to the mean power of the input RF signal, withtemperature, process and supply variation related errors in the peakpower output signal on line 310 cancelled by the subtraction of the peakpower reference output signal on line 398.

As discussed above, it will be recognized by those skilled in the artthat the RF detector, and its various components, may be implemented ina variety of ways, not limited to the above-discussed examples. Forexample, the detector array 340 may be modified to include a series ofreference detectors (optionally, reference squarers) 406, as illustratedin FIG. 18. The array of reference detectors 406 may receive variousbias points 410 from the gain stage 334. The outputs, Iref, from thereference detectors 406 may be fed to summers 408, which receivedetected signals from the detectors 342. The reference signals, Iref,may be subtracted from the detector 342 output signals in the summers408 to cancel process and temperature variations. The output signalsfrom the summers 408 may be fed as signals 352 to the interpolator 344,as discussed above. In one example, one reference detector 406 may beprovided for each detector 342. Alternatively, two or more detectors 342may share the same reference detector 406, resulting in fewer numbers ofreference detectors being used.

Referring to FIG. 19, according to another embodiment, the interpolator344 may be replaced with a coefficient generator 412 which generatesscaling coefficients responsive to the average power output (i.e., thefeedback signal) on line 318. In this example, the detector array 340may include squaring RF detectors 414 which have a variable gain factor.Thus, each detector 414 may output the square of the received inputsignal, adjusted or weighted by the gain factor of the detector. Thescaling coefficients from the coefficient generator 412 may be used toset the gain factors of the detectors 414 in the detector array 340. Theoutputs from the detector array 340 may be summed in summer 424 togenerate the detector output signal on line 306 that is provided to theintegrator 312 (for the average power detection channel) and theinstantaneous power processing block 320. The coefficients may beconfigured so as to “select” (for example, by gain-scaling) one or moreof the detector 414 outputs, similar to the technique discussed abovewith reference to the interpolator. For example, the gain coefficientsmay be configured so as to select the outputs from those detectors 414operating within their optimum square-law region.

According to another embodiment, the variable gain detection subsystem304 may include dual chains of amplifiers/attenuators 418, 420, asillustrated in FIG. 20. The gain taps from each pair ofamplifiers/attenuators (i.e., pairs formed from one amplifier/attenuatorfrom each chain 418, 420) may be multiplied together in multipliers 422which have a variable scale factor and summed in the summer 424 toprovide the signal on line 306. Thus, if the two gain taps forming eachpair are similar, a squaring operation may be achieved by themultiplication. Accordingly, the multipliers 422 may be used in thisembodiment instead of squaring detectors with the multipliers receivingcoefficients generated in the coefficient generator 412. The scalingcoefficients from the coefficient generator 412 may be used to set thescaling factors of the multipliers 422.

In summary, several variations, aspects and embodiments of an RFdetector have been discussed. The RF detector may provide two outputs,one being a function of the true RMS power level of an RF input signal,and the other being a function of the instantaneous/peak power of the RFinput signal, normalized to the average power level. These outputs maybe stable with variations in temperature and supply voltage. The RFdetector may thus optionally also provide a measurement of the crestfactor of the RF input signal, which may be useful in a variety ofapplications. The RF detector may provide accurate measurements of theRF input signal power even in the presence of complex modulationschemes. In one embodiment, by using multiple detectors, and selectingthose detectors operating in their square-law region, the dynamic rangeof the RF detector may be greatly extended up to a maximum level that isdependent on the number of detection stages. Furthermore, because thesame variable gain detection subsystem can be used for both the averagepower measurement and instantaneous/peak power measurement, the same (orvery similar) dynamic range may be achieved for both measurements. Inone example, the RF detector provides an input dynamic range of about 70dB and may provide accurate RMS power measurement over an inputfrequency range of about 100 MHz to 3.9 GHz and over various modulationstandards, including CDMA, TDMA and GSM. The RF detector may provide alinear-in-dB output, which is one embodiment is scaled by 37 millivoltsper dB. The RF detector may be implemented using SiGeCMOS IC processtechnology and may be provided as an integrated circuit in a leadlessSMT package. The following table provides some example specificationsfor one example of an embodiment of the RF detector, measured at roomtemperature:

TABLE 1 Measured Parameter Data Units Input Frequency Range 0.1 to 3.9GHz Input Dynamic Range (to 1 dB measurement error, @  100 MHz 71 dB 900 MHz 70 dB 1900 MHz 70 dB 2700 MHz 66 dB 3500 MHZ 52 dB RSSI Slope(@ 1900 MHz) 37.4 mV/dB RSSI Intercept Input Level (@ 1900 MHz) −68 dBmOutput Error with Temperature +−1 dB Output Deviation for 256 QAMModulation 0.1 dB Instantaneous Power Output Temp. Sensitivity 312 ppm/°C. Instantaneous Power Output Modulation BW 40 MHz (3 dB)* Min. InputReturn Loss (50 Ohm)** 10 dB Bias Voltage (Vcc) - nominal 5 V Min/Maxrange 4.5-55  V Current (Icc) 67-84 mA Operating Temperature Range −40to +85 ° C. *At 900 MHz, −20 dBm RF Power, 4-tone input signal, 3 dBreduction in measured swing. **Measured with balun up to 2.4 GHz. InputReturn Loss is limited by Balun.

Having thus described several aspects of at least one embodiment, it isto be appreciated various alterations, modifications, and improvementswill readily occur to those skilled in the art. Such alterations,modifications, and improvements are intended to be part of thisdisclosure and are intended to be within the scope of the invention.Accordingly, the foregoing description and drawings are by way ofexample only, and the scope of the invention should be determined fromproper construction of the appended claims, and their equivalents.

1. A power detector comprising: an input configured to receive an inputsignal; a variable gain detection subsystem including at least onedetector coupled to the input and which detects the input signal andprovides a detector output signal; an integrator coupled to the variablegain detection subsystem and configured to receive the detector outputsignal and a reference signal and to provide an integrator output signalwhich is representative of an average power level of the input signal;and an instantaneous power processing device coupled to the variablegain detection subsystem and configured to receive the detector outputsignal and to provide at an output of the power detector aninstantaneous power output signal which is representative of theinstantaneous power level of the input signal normalized to the averagepower level of the input signal; wherein the variable gain detectionsubsystem is configured to receive the integrator output signal and toadjust the detector output signal to a level approximately that of thereference signal.
 2. The power detector as claimed in claim 1, whereinthe variable gain detection subsystem includes a variable gain amplifiercoupled between the input and the at least one detector; wherein thevariable gain amplifier is configured to receive the integrator outputsignal and to provide an amplified output signal; wherein the variablegain amplifier is configured so that the gain of the variable gainamplifier is controlled by the integrator output signal; and wherein theat least one detector is configured to receive the amplified outputsignal.
 3. The power detector as claimed in claim 1, wherein the atleast one detector comprises only one squaring detector configured toreceive the input signal; wherein the variable gain detection subsystemfurther includes a bias control circuit coupled between the integratorand the squaring detector; wherein the bias control circuit isconfigured to receive the integrator output signal and to provide adetector gain control signal to the squaring detector; and wherein thebias control circuit is further configured so that scaling of thesquaring detector is controlled by the integrator output signal.
 4. Thepower detector as claimed in claim 1, wherein the variable gaindetection subsystem further includes: at least one reference detectornot coupled to the input and configured to generate a reference signal;and means for subtracting the reference signal from the detector outputsignal; wherein the reference detector is configured to reduce effectsof temperature and fabrication process variations at the output of thedetection subsystem.
 5. The power detector as claimed in claim 1,wherein the variable gain detection subsystem further includes: at leastone reference detector not coupled to the input and configured togenerate a reference signal; and a summer configured to receive thedetector output signal and the reference signal and to output adifference signal based on subtraction of the reference signal from thedetector output signal to reduce effects of temperature and fabricationprocess variations at the output of the detection subsystem.
 6. Thepower detector as claimed in claim 1, where the variable gain detectionsubsystem further includes means for subtraction of a signal generatedby at least one reference detector not coupled to the input; wherein thereference detector is configured to reduce effects of temperature andfabrication process variations at the output of the detection subsystem.7. The power detector as claimed in claim 1, wherein the instantaneouspower processing device comprises: a transistor follower configured toreceive the detector output signal and to provide the instantaneouspower output signal; and a network coupled to the transistor followerand configured to compensate for temperature variation of thebase-emitter voltage of the transistor follower to provide asubstantially temperature stable instantaneous power output signal. 8.The power detector as claimed in claim 7, wherein the network comprisesa plurality of transistors and at least two resistors including a firstresistor and a second resistor; wherein a base of the transistorfollower is coupled to a node between the first resistor and the secondresistor.
 9. The power detector as claimed in claim 8, wherein thetransistors are bipolar transistors.
 10. The power detector as claimedin claim 8, wherein the transistors are field effect transistors. 11.The power detector as claimed in claim 7, wherein the instantaneouspower processing device further comprises a capacitor coupled to thetransistor follower, the capacitor configured to store a voltagerepresentative of a peak signal level at an output of the transistorfollower.
 12. The power detector as claimed in claim 7, wherein theinstantaneous power processing device further comprises: a comparatorconfigured to receive the detector output signal and the referencesignal and to generate an error signal based on a subtraction betweenthe detector output signal and the reference signal; and wherein thetransistor follower is configured to receive the error signal and toprovide the instantaneous power output signal.
 13. A power detectorcomprising: an input configured to receive an input signal; a variablegain detection subsystem including only one squaring detector coupled tothe input and which detects the input signal and provides a detectoroutput signal; an integrator coupled to the variable gain detectionsubsystem and configured to receive the detector output signal and areference signal and to provide an integrator output signal which isrepresentative of an average power level of the input signal; and aninstantaneous power processing device coupled to the variable gaindetection subsystem and configured to receive the detector output signaland to provide at an output of the power detector an instantaneous poweroutput signal which is representative of the instantaneous power levelof the input signal normalized to the average power level of the inputsignal; wherein the variable gain detection subsystem further includes abias control circuit coupled between the integrator and the squaringdetector, the bias control circuit being configured to receive theintegrator output signal and to provide a detector gain control signalto the squaring detector; and wherein the variable gain detectionsubsystem is further configured to control a scaling factor of thesquaring detector based on the integrator output signal to adjust thedetector output signal to a level approximately that of the referencesignal.
 14. A method of power detection comprising: receiving an inputsignal at an input of a detector; detecting with the detector a powerlevel of the input signal to provide a detected signal; subtracting fromthe detected signal a signal not related to the input signal to reduceeffects of temperature and fabrication process variations at an outputof the detector and to provide an output signal; comparing the outputsignal with a reference signal to provide an error signal; integratingthe error signal to provide an integrated signal which is representativeof an average power level of the input signal; providing aninstantaneous power output signal responsive to the output signal, theinstantaneous power output signal being representative of theinstantaneous power level of the input signal normalized to the averagepower level of the input signal; and adjusting the output signal to alevel approximately that of the reference signal.
 15. The method asclaimed in claim 14, wherein adjusting the output signal includesadjusting a scaling factor of the detector based on the integratedsignal.
 16. The method as claimed in claim 14, wherein providing theinstantaneous power output signal includes buffering the output signalusing a transistor follower to provide the instantaneous power outputsignal.
 17. The method as claimed in claim 16, further comprisingcompensating for temperature variation of a base-emitter voltage of thetransistor follower to provide a substantially temperature stableinstantaneous power output signal.
 18. The method as claimed in claim14, wherein subtracting from the detected signal the signal not relatedto the input signal includes subtracting from the detected signal asignal generated by a reference detector which does not receive theinput signal.